The basic converter provides an output
voltage that is about 10 times the input voltage. The voltage boost
occurs when the 2N4401 switches off, and the magnetic flux that had
been supported by current flowing in the inductor, collapses. This
rapid change in flux is accompanied by the voltage on the collector of
the 2N4401 rapidly rising, limited for the most part by the
turnoff time of the
transistor, but ultimately being clamped to a diode drop above the
voltage across the output capacitor. This type of converter is often
referred to as a boost
converter or a flyback converter, the later term referring to the high
voltage power supply that was integrated with the horizontal deflection
circuits in cathode ray tube television sets. High voltages would be
developed when the horizontal scan circuit rapidly made the luminous
spot return (or "fly back") to the left side of the screen.
Oscillation frequency
is a little under 300 kHz when under load. I had first used a 2N2222 as
the NPN because they are pretty common parts, but I could not resist
the improved efficiency of the faster switching 2N4401.
After measuring the input voltage required to maintain the output
voltage at +24 volts as the load was varied, the basic circuit was then
treated as a building block and placed inside an analog regulation
loop that could supply the range of voltages needed. From the graph
(further below) ,
the analog regulator's output, the emitter of the 2N2222, needs to be
able to cover the range from 2.2 volts to about 3.5 volts in order to
supply +24 Volts from no load to a 5 milliamp load.
After some refinement to the basic
converter, I determined that the converter's analog section should
also be able to provide about 40 milliamps to the converter section
when supplying a load current of 5 milliamps.
The completed circuit is shown
above. The analog regulator is a TL431 shunt regulator chip. It
is followed by Q1, a
2N2222 which both provides more output current, and shifts the voltage
one base-emitter drop lower. The TL431 adjusts the current through its
cathode, and thus the 500 Ohm resistor to
maintain 2.5 volts on the reference input. The voltage divider on the
output of the supply divides the +24 volts down to +2.5 volts. I added
a pot to adjust the output voltage since the build up of the tolerances
could result in more than a volt of error.
The voltage at
the cathode of the TL431 is a function of the current drawn by the
TL431, and this voltage is buffered by the 2N2222. The 10 Ohm resistor
in series with the 2N2222 limits the peak emitter current that is
available to change C1 during turn-on and in case there are large
swings in the load current. C4 and R10 reduce the closed loop gain at
high frequencies to keep the TL431 from oscillating.
(Above) Input voltage, in volts,
applied to the converter stage to
maintain +24 volts at the output
varied
as a function of the load current in amps. Note that for this test, the
input to the regulator circuit was +9VDC. When powered from a +5VDC
supply, the maximum load current before dropping out of regulation is
6.8 milliamps.
(Above)
Input current, in amps,
drawn by the converter stage while maintaining +24 VDC on the output,
as a function of load, in amps. The current determined by measuring the
voltage drop across the 10 Ohm resistor in series with the collector of
the 2N2222.
Since the TL431 cathode is designed to operate at +2.5 VDC, the base
emitter drop of the 2N2222 assured that the voltage supplied to the
converter could be lower than 2.0 volts, to accommodate no load. The
value of R1 was chosen so as to provide the minimum required current
through the TL431 when operating at the maximum output voltage, and
still provide sufficient base current to supply over 40 milliamps of
emitter current.
A casual glance will show some differences between the basic converter
stage and the one in the completed circuit.
The addition of D2 prevents the flyback pulse from avalanching the
base-emitter junction with the collector pulse from Q3. This might have
degraded the beta of Q2 over the long run. A significant amount of
power was also being dissipated in the avalanched base-emitter
junction. R4 helps Q2 turn off more quickly than
it would if R4 were not in the circuit.
A modification to the base drive circuit for Q3 helps Q3 turn off more
quickly, thus saving more power. When Q3 is driven by current from Q2
through R3, inductor L1 draws some of that base current. When Q2 turns
off, the current in L1 causes the base of Q3 to swing negative, thus
turning it off more quickly. R6, the 330 Ohm resistor across L1,
absorbs some of the energy from L1 so that the voltage on the base of
Q3 does not ring positive during the flyback pulse, thus saving several
milliamps of input current. In a higher power circuit, Q3 turning
on during the flyback pulse might well mean destruction of Q3.
The base drive enhancements are not required, as they only add to the
efficiency of the circuit. D2, R4, D1, and L1 can all be omitted, and
the original base drive circuits shown in the basic converter circuit
can be used instead, only the current drain will be a lot higher. The
input voltage and input current curves on this
web page were measured on the completed, improved circuit, inside the
regulation loop.
The value for C3 was calculated to limit the maximum positive-going
voltage transient on the output in case the load became disconnected.
If all of
the energy stored in L2 was suddenly dumped into C3, how large would C3
have to be to keep the output voltage from rising more than 1 volt?
A safe and practical answer is found by quickly by ignoring the
resistance of the
circuit and noting that the energy stored in an inductor is 1/2 *
L*I^2, and the energy stored in a capacitor is 1/2 * C * V^2. Using 220
microhenries and taking the inductor's saturation current of 300
milliamps as the maximum current, C = (L*I^2)/V^2 = 20 uf.